Signal processing systems and signal processing methods

ABSTRACT

It is provided a signal processing system, comprising at least a first, a second and a third digital-to-analog converter (DAC); a processing unit configured for splitting a sampled signal into a first and a second signal corresponding to different frequency portions of the sampled signal, transmitting the first signal to the first DAC, splitting the second signal into a first and a second subsignal and transmitting the first subsignal to the second DAC and the second subsignal to the third DAC, the first subsignal corresponding to the real part of the second signal and the second subsignal corresponding to the imaginary part of the second signal; an IQ mixer configured for mixing an analog output signal of the second DAC and an analog output signal of the third DAC and a combiner for combining an analog output signal of the first DAC and an output signal of the IQ mixer.

CROSS-REFERENCE TO A RELATED APPLICATION

This application is a National Phase patent application of InternationalPatent Application Number PCT/EP2015/077000, filed on Nov. 18, 2015.

BACKGROUND

The invention relates to signal processing systems and to a signalprocessing method.

The increasing need for high data rates raises the requirements onhigh-speed communication systems. For today's communication systemsflexible transmitters based on digital-to-analog converters (DACs) aredesirable. They are able to vary the modulated bandwidth and themodulation format of the transmitted signal. The performance of thesedevices is determined by their bandwidth and their sample rate.

However, analog components of a DAC impose limitations on the totalbandwidth achievable by such a device such that systems using multipleparallelized (interleaved) DACs have been developed. Interleaving theDAC output signals may be carried out in the time or in the frequencyregime. For example, U.S. Pat. No. 7,535,394 describes a hardware setupfor frequency interleaving. However, the quality of the output signal ofthe system may be restricted due to system impairments.

SUMMARY

The objective of the invention is to improve the quality, the samplerate and/or the bandwidth of the output signal of a system consisting ofmultiple DACs.

According to the invention, a signal processing system is provided,whereby the system comprises

-   -   at least a first, a second and a third digital-to-analog        converter (DAC);    -   a processing unit configured for splitting a sampled signal into        a first and a second signal corresponding to different frequency        portions of the sampled signal, transmitting the first signal to        the first DAC, splitting the second signal into a first and a        second subsignal and transmitting the first sub signal to the        second DAC and the second sub signal to the third DAC, the first        sub signal corresponding to the real part of the second signal        and the second sub signal corresponding to the imaginary part of        the second signal;    -   an IQ mixer configured for mixing an analog output signal of the        second DAC and an analog output signal of the third DAC;    -   a combiner for combining an analog output signal of the first        DAC and an output signal of the IQ mixer.

The IQ mixer permits to use three DACs for generating the desired outputspectrum, which might have a three times larger analog bandwidthcompared to a single DAC. This increase in bandwidth might be achievedwithout introducing significant computational overhead. The first DAChandles a first spectral portion of the sampled signal, while the IQmixer and thus the second and the third DAC handle a second and a thirdspectral portion of the sampled signal.

Due to the IQ mixer the spectrum of the second signal does not need tohave conjugate symmetry properties (corresponding to a real valued timedomain signal). Thus, the spectrum can be defined for positive as wellas for negative frequencies, wherein the resulting time domain signalmay be complex valued.

The processing unit may be configured for carrying out the splitting ofthe sampled signal into the first and the second signal in the frequencydomain. In particular, the first spectral portion of the sampled signalcorresponds to a real valued signal (the first signal) which is feddirectly to the first DAC. The second spectral portion may not showconjugate symmetry such that the corresponding time domain signal iscomplex. For example, after using a Fourier transform (e.g. an IFFT) onthe second signal for generating a time domain signal, this time domainsignal is separated into the real and the imaginary part (the first andthe second subsignal), which are supplied to the second and third DAC,respectively. Accordingly, the processing unit might be configured forcarrying out the splitting of the sampled signal into the first and thesecond signal in the time domain and/or may be configured for carryingout a Fourier transform of the second signal for generating the firstand the second subsignal.

Splitting of the non-conjugate-symmetrical second signal spectrum forgenerating the first and the second subsignal supplied to the second andthe third DAC can be performed in the spectral domain instead of thetime domain as well. Accordingly, the processing unit may be configuredfor carrying out the splitting of the sampled signal into the first andthe second signal in the frequency domain, wherein the processing unitmay also be configured for carrying out a Fourier transform of thesecond signal for generating the first and the second subsignal. Forexample, the individual signals for inphase (the first subsignalsupplied to the second DAC) and quadrature (the second sub signalsupplied to the third DAC) can be obtained each by an IFFT of thecorresponding spectral components directly. This variant may require theexploitation of general symmetry properties of the Fouriertransformation for odd and even functions and spectra, respectively.

It is further noted that, of course, more than three DACs and/or aplurality of IQ mixers could be used. Moreover, the DACs do not have tobe standard single DACs. For example, the first, second and/or third DACmay be realized by a plurality of sub-DACs. For example, at least one ofthe DACs is implemented by a TIDAC arrangement using time interleaving,e.g. digital time interleaving in combination with an analog summationpoint or a MUXDAC arrangement comprising an analog multiplexer (seedetailed explanation below).

The frequency portion of the sampled signal that corresponds to thefirst signal may comprise lower frequencies than the frequency portionof the sampled signal that corresponds to the second signal.

According to another embodiment of the invention, the system may furthercomprise a low pass filter for filtering the outputs of the DACs and/ora band pass filter for filtering the output of the IQ mixer. Thesefilters, however, are only optional. It is also possible that no analogfilters are used or that at least the low pass filter or the band passfilter is omitted.

Further, the processing unit may be realized by a digital signalprocessor (e.g. in the form of a programmed device, i.e. a programmabledevice equipped with a corresponding software).

The combiner might be a passive combiner such as e.g. a power combiner,a (e.g. frequency selective) diplexer or triplexer. Further, thecombiner may be an active device (e.g. a summing or a differentialamplifier).

Moreover, the IQ mixer (IQ modulator) might be configured for singlesideband modulation.

For example, the IQ mixer is an electronic device. However, the IQmodulator may also be realized by an opto-electronic modulator. It isnoted that the invention is of course not restricted to a system usingthree DACs and/or one IQ mixer. Rather, more than three DACs and morethan one IQ mixer (e.g. more than one IQ modulator or combinations ofmodulators) could be provided. In particular, combinations of differentmodulators could be used, e.g. combinations of at least one IQmodulator, at least one RF modulator and/or at least a modulator forsingle sideband (SSB) modulation. For example, eight DACs (DAC 1-8)might be used, wherein DAC 1 processes the baseband, DAC 2 supplies theinphase component of a signal to a first IQ mixer, DAC 3 supplies thequadrature component of the signal to the first IQ mixer, DAC 4 suppliesa signal to a first RF modulator, DAC 5 supplies a signal to a SSBmodulator, DAC 6 supplies a signal to a second RF modulator, DAC 7supplies the inphase component of a signal to a second IQ mixer and/orDAC 8 supplies the quadrature component of the signal to the second IQmixer. Of course, only some of these components or additional componentsmight be used.

The invention in a second aspect relates to a signal processing method,in particular using the system described above, the method comprisingthe steps of:

-   -   providing at least a first and a second digital-to-analog        converter (DAC);    -   splitting a sampled signal into at least a first and a second        signal corresponding to different frequency portions of the        sampled signal by means of a processing unit;    -   pre-equalizing the first and the second signal;    -   converting the pre-equalized first signal into a first analog        signal using the first DAC;    -   converting the pre-equalized second signal into a second analog        signal using the second DAC;    -   combining the first and the second analog signal using a        combiner,    -   wherein the processing unit, the first DAC and the combiner        define a first processing channel,    -   wherein the processing unit, the second DAC and the combiner        define a second processing channel, wherein    -   the pre-equalized first signal is generated by processing the        first signal in such a way that the pre-equalized first signal        compensates cross talk between the first and the second        processing channel, and/or the pre-equalized second signal is        generated by processing the second signal in such a way that the        pre-equalized second signal compensates cross talk between the        first and the second processing channel.

Thus, the method according to the invention generally relates toincreasing the bandwidth by splitting the sampled input signal into atleast a first and a second signal, using at least two DACs fordigital-to-analog conversion and recombining the analog signals (“analogbandwidth interleaving”—ABI). Meaning that, instead of using a singleDAC, a plurality of DACs is used, wherein, in principle, an arbitrarynumber of DACs could be used. The quality of the analog output signal isimproved by taking into account cross talk effects between multipleprocessing channels (at least 2) when generating the pre-equalizedsignals. The generation of the pre-equalized first and the second signalmight be performed by digital signal processing (e.g. by means of theprocessing unit or another digital signal processor).

It is noted that the first and the second processing channel mightcomprise further components such as analog filters. These furthercomponents may be arranged between the processing unit and the combiner.However, it is also possible that further components (e.g. a filter) forprocessing the output of the combiner (i.e. components arranged behindthe combiner) form part of the first and the second processing channel,respectively, and are taken into account when generating thepre-equalized signals.

According to an embodiment of the invention, generating thepre-equalized first and second signal is carried out using the resultsof a calibration measurement with respect to at least a (e.g. spatial,frequency and/or time portion) of the first and/or the second processingchannel.

In order to compensate the analog impairments of e.g. a mixer, filters,couplers, the combiner and/or the frequency response of the DACs,information about the frequency responses of these components is needed.The calibration measurement may use a channel estimation algorithm toretrieve this information for the whole system. Further, the calibrationmight be carried out using an external receiver or an internal receiver(that might form a common unit with the DACs).

For example, the calibration measurement is carried out using a channelestimation scheme with respect to the first and/or the second processingchannel. It is noted that specific sequences for the channel estimationare not absolutely necessary, but might improve the channel estimationquality. According to an embodiment of the invention, a first channelestimation sequence is transmitted to the first DAC (and e.g. via thefirst processing channel) and a second channel estimation sequence istransmitted to the second DAC (and e.g. via the second processingchannel), wherein the first channel estimation sequence isdistinguishable from the second channel estimation sequence. Forexample, orthogonal sequences might be transmitted over the first andthe second processing channel for channel estimation (the sequencesbeing e.g. orthogonal with respect to frequency, time, code and/orspatially).

It is also possible that the calibration measurement comprises an S-and/or X-parameter measurement of at least a part of the analog sectionof the first and/or the second processing channel. For example, usingthe S- and/or X-parameter measurement results information about signalimpairments induced by components of the multiple DAC system can beobtained. The system can be either measured as a whole or thecomponent's parameters are measured individually and are digitallycombined afterwards, wherein the measurement results can be used forfactory calibration of at least some of the components. During theoperation of the DAC system, the system might need to compensate achange of parameters, e.g. temperature variations of the components etc.

Accordingly, the pre-equalized first and second signal may be generatedadaptively by means of the results of re-calibration measurementscarried out using a portion of an analog signal produced by thecombiner. That is, pre-equalization of the first and second signal maybe adapted continuously during the operation of the first and the secondDAC. The re-calibration might comprise measuring distinct frequencylines or frequency ranges. Further, the re-calibration may includetracking the local oscillator of a mixer (e.g. the IQ mixer describedabove) of the system and the compensation of phase and/or frequencydeviations.

It is further noted that, of course, the IQ mixer system described abovemight also use the pre-equalizing method. That is, a pre-equalized firstand second signal (and thus a pre-equalized first and second subsignal)might be generated and supplied to the first, the second and the thirdDAC. For example, the pre-equalized signals are generated (based on thefirst and the second signal) in such a way that they compensate crosstalk between the processing channels (formed between the processingunit, the DACs and the combiner) and/or compensate an IQ imbalance ofthe IQ mixer (i.e. cross talk between the I and Q path of the IQ mixer).

For example, the calibration measurement comprises at least one of thefollowing steps:

-   -   generating a channel estimation sequence (CE sequence);    -   running the channel estimation sequence with the first DAC;    -   acquiring the analog signal produced by the first DAC;    -   running the channel estimation sequence with the second DAC;    -   acquiring the analog signal produced by the second DAC;    -   estimating a path delay between the first and the second        processing channel, e.g. with cross-correlation;    -   performing the channel estimation using a channel estimation        scheme;    -   load calibration data obtained by the channel estimation (e.g.        in the form of a data sequence) into a memory for use by the        processing unit;    -   pre-compensating the first and the second signal (sequence)        using the calibration data in the frequency domain and        transforming the (two) spectra to the time domain (e.g. by means        of an IFFT);    -   running the pre-compensated first and second sequence with the        first and the second DAC, respectively.

It is noted that the first and the second DAC may be synchronized.Moreover, it is noted that the channel estimation sequence may besupplied to the first and the second DAC simultaneously.

Further, according to another embodiment of the invention, the channelestimation scheme comprises treating the combination of the first andthe second processing channel as a MIMO (multiple input multiple output)system. For example, the calibration measurement comprises determiningcoefficients of a frequency response matrix related to the MIMO system.If the first and the second processing channel is treated as a MIMOsystem, the cross coupling between the channels can be modeled as alinear system in the frequency domain. However, the description as astandard MIMO problem may fail as cross talk components may be reversedin frequency. Hence, a modified MIMO formulation might be used as willbe explained by means of the following exemplary formulation.

DSP (Digital Signal Processing) Operations

The sampled input signal may be a 2N-point data sequence d=[d(0), d(1),. . . , d(2N−1)] ∈

has to be split in the Fourier domain in order to apply it to the twoDACs.

The corresponding spectrum D=

(d)=[D⁺,D⁻] (Fourier transformation) is split into first and secondspectra D₁ and D₂:D=[D ⁺ ,D ⁻]=[D ₁ ⁺ ,D ₂ ⁺ ,D ₂ ⁻ ,D ₁ ⁻]D ₁=[D ₁ ⁺ ,D ₁ ⁻]D ₂=[D ₂ ⁺ ,D ₂ ⁻],where the plus and minus superscript denote the positive and thenegative frequency range of the spectrum, respectively. It is noted thatthe input signal does not have to be a time domain signal. Rather, if afrequency domain modulation format, e.g. DMT or OFDM, is utilized, theinput signal is first given by its frequency domain representation andmight be directly split. Thus, arbitrary signals in both time andfrequency domain can be processed. The splitting operation maycorrespond to a brick-wall filtering of a lower sideband and both abrick-wall filtering and implicit downconversion of the upper sideband.A digital local oscillator for downmixing the higher sideband is notmandatory. Further, other splitting approaches such as raised cosinefiltering are possible as well, but they may introduce overhead and thusmay reduce the achievable overall bandwidth. Though, they might improvethe time domain behavior, since the impulse response is limited inlength.

Having obtained the two data spectra (the first and the second signal),pre-equalization is performed according toX ₁ =f _(EQ1)(D ₁ ,D ₂)X ₂ =f _(EQ2)(D ₁ ,D ₂),where f_(EQi)O, with i∈{1,2} represents an arbitrary function performingequalization. The sequences x_(i)=

⁻¹{X_(i)} (inverse Fourier transformation) with i∈{1,2} are supplied tothe first and the second DAC, respectively.

It is noted that the following explanations regarding the definition ofthe processing channels provide that the pre-equalizer is inactive suchthat X₁(k)=D₁(k) and X₂(k)=D₂(k),

Obtaining a Channel Model

If using a pre-equalizer to remove distortions, the input spectrum canbe altered in the frequency region [0,f_(s)/2] by the first DAC and inthe region [f_(s)/2,f_(s)] by the second DAC due to an upconversion.Frequency components located at frequencies f>f_(s) cannot be directlymodified. However, the undesired components at f>f_(s) can be shaped byan appropriate filter or removed by utilizing oversampling incombination with an appropriate steep roll-off filter. This opens thepossibility to apply DSP to undo cross-talk effects between the firstand the second processing channel.

The received signal s(n) (the output analog signal) is restricted to thefrequency range [−f_(s),f_(s)], corresponding to a sampling rate of2f_(s) and a spectral representation S. The spectrum is split into alower frequency band and an upper frequency bandS=[S ⁺ ,S ⁻]=[S _(I) ⁺ ,S _(II) ⁺ ,S _(II) ⁻ ,S _(I) ⁻].S _(I)=[S _(I) ⁺ ,S _(I) ⁻]S _(II)=[S _(II) ⁺ ,S _(II) ⁻]yielding two signal spectra that are individually modifiable by theDACs.MIMO Problem Identification

In the following, the individual frequency components of the spectrabecome important. The notation focuses on these by referring to S(k)instead of S, whereby k denotes the discrete frequency.

For a solution of the problem the transformation of D₁ (k) and D₂ (k) toX₁(k) and X₂(k) has to be calculated. The actual problem encompassescross coupling terms which are the mirror spectra of the actual spectra,which are reversed in frequency and complex conjugated. This can bedescribed by a shift of half of the sample points of the DFT:X(k+N/2)=X(k−N/2), exploiting the repetitive nature of the discretespectrum.

The cross-coupling problem, which arises due to the non-ideal filteringis denoted asS _(I)(k)=H _(II)(k)·X ₁(k)+H ₁₂(k)·X ₂(k+N/2)S _(II)(k)=H ₂₁(k)·X ₁(k+N/2)+H ₂₂(k)·X ₂(k),where X₁ (k) and X₂(k) are the spectra after the equalizer (i.e. aftergenerating the first and the second pre-equalized signal).

The derived model is a special 2×2 MIMO model and can be rewritten as astandard 4×4 MIMO model as shown below. Other approaches as a 1×1 or a2×1 or even a 4×1 model are possible as well.

MIMO Problem Solution

As set forth above, the 2×2 MIMO ABI model is a linear equation systemgiven byS _(I)(k)=H _(II)(k)·X ₁(k)+H ₁₂(k)·X ₂(k+N/2)S _(II)(k)=H ₂₁(k)·X ₁(k+N/2)+H ₂₂(k)·X ₂(k),

Recovering the original spectrum requires a transformation of D₁(k) andD₂ (k) to X₁(k) and X₂(k), which reverts the MIMO channel effects asexplained above. Therefore, the following conditions have to be met forS_(I) (k) and S_(II)(k):S _(I)(k)

D ₁(k)S _(II)(k)

D ₂(k).

The above linear equation system can be solved, e.g. by solving thefirst equation for X₁(k) in terms of X₂ (k+N/2) and then substitutingthis expression into the second equation.

In general, the MIMO system is described in the frequency domain asS(k)= C (k)X(k)+V(k),whereby there is an individual equation for each frequency line k. S(k)is the output signal, X(k) is the DAC output signal and V(k) is thevector of noise samples. These are given by

${{S(k)} = \begin{bmatrix}{S_{1}(k)} \\{S_{1}( {k + {N/2}} )} \\\vdots \\{S_{L}(k)} \\{S_{L}( {k + {N/2}} )}\end{bmatrix}},{{X(k)} = \begin{bmatrix}{X_{1}(k)} \\{X_{1}( {k + {N/2}} )} \\\vdots \\{X_{M}(k)} \\{X_{M}( {k + {N/2}} )}\end{bmatrix}},{{V(k)} = \begin{bmatrix}{V_{1}(k)} \\{V_{1}( {k + {N/2}} )} \\\vdots \\{V_{L}(k)} \\{V_{L}( {k + {N/2}} )}\end{bmatrix}},$whereby L and M denote the number of receivers (e.g. frequency bands atDAC system output) and transmitters (e.g. frequency bands aftersplitting), respectively. The channel matrix C(k) is defined as

${\underset{\_}{C}(k)} = \begin{bmatrix}{C_{1,1}(k)} & {C_{1,2}(k)} & \ldots & {C_{1,{2M}}(k)} \\{C_{2,1}(k)} & {C_{2,2}(k)} & \ldots & {C_{2,{2M}}(k)} \\\vdots & \vdots & \ddots & \vdots \\{C_{{2L},1}(k)} & {C_{{2L},2}(k)} & \ldots & {C_{{2L},{2M}}(k)}\end{bmatrix}$

The pre-equalizer undoing the channel impairments is given by theweight-matrix W(k):X(k)= W (k)D(k),whereby W(k) and D(k) are given by

${{\underset{\_}{W}(k)} = \begin{bmatrix}{W_{1,1}(k)} & {W_{1,2}(k)} & \ldots & {W_{1,{2L}}(k)} \\{W_{2,1}(k)} & {W_{2,2}(k)} & \ldots & {W_{2,{2L}}(k)} \\\vdots & \vdots & \ddots & \vdots \\{W_{{2M},1}(k)} & {W_{{2M},2}(k)} & \ldots & {W_{{2M},{2L}}(k)}\end{bmatrix}},{{D(k)} = {\begin{bmatrix}{D_{1}(k)} \\{D_{1}( {k + {N/2}} )} \\\vdots \\{D_{L}(k)} \\{D_{L}( {k + {N/2}} )}\end{bmatrix}.}}$

There is a plurality of suited equalizers for obtaining X(k) such aszero forcing equalizers, minimum mean square error (MMSE) equalizers oradaptive equalizers. As adaptive equalizer e.g. a least mean squareequalizer might be used, wherein, however, other types such as recursiveleast squares (RLS) equalizers are possible as well as nonlinearadaptive equalizers. Using a least mean square (LMS) equalizer foradaptive equalization, the equalizer coefficients for the LMS areupdated according toW _(i+)(k)= W _(i)(k)+μ(D _(i)(k)−S _(i)(k))X _(i)*(k)with μ being the update coefficient of the LMS algorithm. By using thisequalizer only the frequency components in the spectral range [0, f_(s)]can be altered. Frequency components f>f_(s) are influenced indirectlyby modifying the frequency components close to f<f_(s) Oversampling (seebelow) can be incorporated to shift the image bands to higher frequencyregions. Hence, cancellation of the images is possible by applying asmall oversampling factor to the data, e.g. 10%, in combination with agood (i.e. steep roll-off) analog filter. For continuously adapting thesystem, the state of the system might be continuously tracked. For this,e.g. a sampling oscilloscope could be inserted in the system forpermitting recalibration during system operation.

It is noted that all operations carried out in the frequency domain canbe of course performed analogously in the time domain. Further, for acontinuous data stream additional known techniques for frequency domainequalization, such as overlap-add or overlap-save, might be introduced.Another possibility is the realization of a time domain equalizerenabling continuous operation.

The invention also relates to a signal processing system, in particularfor carrying out the method described above, the system comprising:

-   -   a processing unit configured for splitting a sampled signal into        at least a first and a second signal corresponding to different        frequency portions of the sampled signal;    -   a pre-equalizing unit for pre-equalizing the first and the        second signal;    -   at least a first digital-to-analog converter (DAC) for        converting the pre-equalized first signal into a first analog        signal and a second DAC for converting the pre-equalized second        signal into a second analog signal;    -   a combiner for combining the first and the second analog signal,    -   wherein the processing unit, the first DAC and the combiner        define a first processing channel,    -   wherein the processing unit, the second DAC and the combiner        define a second processing channel,    -   wherein the pre-equalizing unit (that may be part of the        processing unit) is configured for generating the pre-equalized        first signal by processing the first signal in such a way that        the pre-equalized first signal compensates cross talk between        the first and the second processing channel, and/or for        generating the pre-equalized second signal by processing the        second signal in such a way that the pre-equalized second signal        compensates cross talk between the first and the second        processing channel.

A further signal processing method comprising the steps of:

-   -   providing at least a first and a second digital-to-analog        converter (DAC);    -   splitting a sampled signal into a first and a second signal        corresponding to different frequency portions of the sampled        signal using a processing unit;    -   creating a first analog signal based on the first signal using        the first DAC;    -   creating a second analog signal based on the second signal using        the second DAC;    -   combining the first and the second analog signal using a        combiner, and    -   creating an oversampled first signal and converting the        oversampled first signal by the first DAC in order to obtain the        first analog signal and/or creating an oversampled second signal        and converting the oversampled second signal by the second DAC        in order to obtain the second analog signal.

For example, the oversampled first signal is created by oversampling thesampled signal and/or the first signal, and/or the oversampled secondsignal is created by oversampling the sampled signal and/or the secondsignal.

The oversampling in the sampled signals (the digital spectra) is used inorder to move the image bands (of the analog output signal) away fromthe desired bands. Thus, appropriate analog filters could be used toeliminate the image bands (e.g. almost completely) and/or to avoid crosstalk. However, the oversampling method might be combined with thepre-equalizing method described above, i.e. the first and the secondsignal are pre-equalized before being supplied to the DACs.

The oversampling might be carried out by inserting zeros in the spectrumof the sampled signal, the first signal and/or the second signalcorresponding to a sinc-interpolation in the time domain. Further, theoversampling may be carried out by raised cosine filtering in thespectrum of the sampled signal, the first signal and/or the secondsignal. It is further noted, that other oversampling approaches arefeasible as well.

The input spectrum (the sampled signal) might be divided unequally amongthe at least two DACs, since the signal in a first processing path(including the first DAC) undergoes filtering only once and thus doesnot necessarily require oversampling both at high frequencies and at lowfrequencies. The second signal may not be generated at base band, but atan intermediate frequency. This process is called digital upmixing ordigital upconversion. Thereby, oversampling by inserting zeros might beapplied. Then, the signal is upconverted with a digital local oscillatorto the desired frequency. A digital image rejection filter may removethe undesired side band. Thus, spectral zeros may be generated both athigh and low frequencies.

When combining the first and the second analog signal, cross talk termsdistorting the signal may not be present. Only the characteristics ofthe analog filters and the DAC sinc-roll-off might have to becompensated.

The resulting ABI problem may be described asS _(I)(k)=H ₁₁(k)·X ₁(k)S _(II)(k)=H ₂₂(k)·X ₂(k)

Thus, the above described MIMO processing may not be needed and theprocessing channels (including the first and the second DAC) might beequalized using two single input single output (SISO) filters.

Another variant of the invention uses oversampling solely either for thefirst DAC or for the second DAC. In the exemplary case of usingoversampling for the first DAC only, the MIMO problem (see above) can bereduced to

${X_{1}(k)} = {\frac{1}{H_{11}(k)} \cdot ( {{D_{1}(k)} - {\frac{H_{12}(k)}{H_{22}( {k + {N/2}} )} \cdot {D_{2}( {k + {N/2}} )}}} )}$${X_{2}(k)} = {\frac{1}{H_{22}(k)} \cdot {{D_{2}(k)}.}}$

Thus, the need for equalization is reduced compared to the case withoutoversampling whereby being able to run at a higher overall bandwidthcompared to the above described second aspect of the invention. Thedescribed oversampling method may be used with the IQ system discussedabove.

The invention further is related to a signal processing system, inparticular for carrying out the method described above, the systemcomprising:

-   -   a processing unit for splitting a sampled signal into a first        and a second signal corresponding to different frequency        portions of the sampled signal;    -   at least a first digital-to-analog converter (DAC) for creating        a first analog signal based on the first signal and a second DAC        for creating a second analog signal based on the second signal;    -   a combiner for combining the first and the second analog signal;        and    -   an oversampling unit for creating an oversampled first signal,        wherein the first DAC is configured for converting the        oversampled first signal in order to obtain the first analog        signal and/or for creating an oversampled second signal, wherein        the second DAC is configured for converting the oversampled        second signal by the second DAC in order to obtain the second        analog signal.

BRIEF DESCRIPTION OF THE DRAWINGS

Embodiments of the invention are described hereinafter with reference tothe drawings.

FIG. 1 shows a block diagram of a signal processing system according toan embodiment of the invention.

FIG. 2 shows a modification of FIG. 1.

FIG. 3 shows a diagram illustrating an embodiment of a signal processingmethod according to the invention.

FIG. 4 shows a block diagram of the system of FIG. 1 showing spectra atspecific positions of the system.

FIG. 5 shows a diagram illustrating portions of the received frequencyspectra.

FIG. 6 shows a diagram illustrating components of the received frequencyspectra.

FIG. 7 shows a block diagram illustrating the MIMO model.

FIG. 8 shows a block diagram of a signal processing system according toanother embodiment of the invention.

FIG. 9 shows a modification of FIG. 8.

FIG. 10 shows a diagram illustrating the spectral splitting scheme usingthe system of FIG. 8 or 9.

FIG. 11 shows possible implementations of the DACs.

FIG. 12 shows a block diagram illustrating a calibration setup.

FIG. 13 shows different realizations of a channel identification system.

FIG. 14 shows a diagram illustrating channel estimation schemes.

FIG. 15 shows a modification of FIG. 14.

FIG. 16 shows another modification the channel estimation scheme.

FIG. 17 shows a diagram illustrating MIMO digital signal processing.

FIG. 18 shows a diagram further illustrating MIMO digital signalprocessing.

FIG. 19 shows the concept of raised cosine filtering.

FIG. 20 shows the frequency response of raised cosine frequency domainfilters and brickwall filters, respectively.

FIG. 21 shows a diagram illustrating another embodiment of a signalprocessing method according to the invention.

FIG. 22 shows a block diagram of the system of FIG. 1 showing spectra atspecific positions of the system when carrying out the methodillustrated in FIG. 21.

DETAILED DESCRIPTION

FIG. 1 depicts a signal processing system 1 according to an embodimentof the invention. The system 1 comprises a processing unit 21 and adigital signal processor 22. Further, the system comprises a first and asecond digital-to-analog converter (DAC) 31, 32 and a combiner 4.

The processing unit 21 receives a sampled input signal d(n) and splitsthe sampled signal d(n) into a first and a second signal d₁(n), d₂(n)corresponding to different frequency portions of the sampled signal. Thedigital signal processor 22 realizes a pre-equalizing unit 220generating a pre-equalized first signal x₁(n) and a pre-equalized secondsignal x₂(n) by processing the first signal d₁(n) and the second signald₂(n) preferably jointly. The pre-equalized first and second signalsx₁(n), x₂(n) are converted into a first and a second analog signal s₁(t)and s₂(t) by means of the first and the second DAC 31, 32, respectively.Of course, the processing unit 21 might also be realized by the digitalsignal processor 22.

After analog filtering (using filters 51, 52, 53) and upmixing thesecond analog signal s₂(t) (using a mixer 6 comprising a localoscillator 61) the final analog signals s₁′(t) and s₂′″(t) are created.The finalized analog signals s₁′(t) and s₂′″(t) are combined with thecombiner 4 to produce the combined output signal s(t). The operation ofthe processing system 1 is also illustrated in FIG. 3.

The processing unit 21, the first DAC 31 and the combiner 4 define afirst processing channel 101, while the processing unit 21, the secondDAC 32 and the combiner 4 define a second processing channel 102. It isnoted that the filters 51, 52, 53 might be part of the processingchannels 101, 102 as well. The digital signal processor 22 generates thepre-equalized first signal x₁(n) in such a way that it compensates crosstalk between the first and the second processing channel 101, 102,and/or generates the pre-equalized second signal x₂(n) in such a waythat it compensates cross talk between the first and the secondprocessing channel 101, 102. Details of the cross talk compensation havebeen discussed above.

It is noted that at least one of the analog filters 51, 52, 53 can beomitted as shown in FIG. 2. It is also possible that no analog filtersare employed at all. However, a low pass filter 50 might be insertedafter combining the signals (i.e. after the combiner 4) in order tosuppress the upper sideband of the mixer 6. The filter 50 could beplaced behind the mixer 6 as well. If the number of DACs exceeds two(and thus more than one mixer is used), the filter 50 is used forsuppressing the sideband of the mixer with the highest LO frequency.

An overview of an embodiment of the method according to the invention isillustrated in FIG. 3 which might be carried out using the system shownin FIG. 1 or 2. A 2N-point digital input signal (sequence), whichcorresponds to the input signal d(n) in FIGS. 1 and 2, undergoes adiscrete Fourier transformation (DFT) to obtain the spectralrepresentation of the sequence (shown in step 1), wherein I⁺, I⁻, II⁺and II⁻ describe the positive and the negative frequency bands of thesignal portions I and II, respectively. The arrows denote the directionof the frequency of the bands.

As illustrated in step 2, the input signal is split into two portions ofequal length N(2) in the spectral domain (e.g. using the processing unit21 of FIG. 1 or 2). These two spectral (frequency) portions are theneach transformed to the time domain by an inverse discrete Fouriertransformation (IDFT). In the next step, the digital signals are fed tothe DACs (e.g. DACs 31, 32 of FIG. 1 or 2), which, for example, arerunning at their maximum sample rate fs without any oversampling inorder to generate the analog signals (step 3).

Due to the zero-order-hold (ZOH) operation of the DACs, the DAC outputsignals are attenuated by a sinc-function, which is indicated bytriangles in step 3. The image bands are removed by appropriate low passfilters (step 4). However, the filters will not filter all image bandcomponents due to their finite roll-off characteristics. Analogprocessing of the first analog signal in the first processing path (seeprocessing path 101 in FIG. 1 or FIG. 2) is finished, while the secondanalog signal undergoes further processing steps. For example, thesecond analog signal is upmixed (e.g. multiplied by a local oscillator(LO) such as the LO 61 in FIG. 1 or 2) in order to be shifted to ahigher frequency region (step 5).

For example, two alternatives 1 and 2 exist for the frequency positionof the LO. Either the LO is located at half of the sampling frequencyfs/2 or the LO is located directly at the sampling frequency fs. Forcarrying out the second alternative, the corresponding spectrum has tobe digitally inverted prior to the D/A conversion in order to ensure theright frequency orientation in the upper band at the end of theprocessing. The upconversion with a cosine carrier will generate twoside bands. One of these side bands is redundant and can be removed by aband pass filter (step 6). Finally, the two individual analog signalsare combined (using e.g. the combiner 4 in FIG. 1 or 2) in order to forman analog representation of the digital input waveform with a bandwidthof fs and a sampling rate of 2 fs (step 7).

In case the mixer 6 shall be omitted, Beyond-Nyquist signaling can beused: The second signal would be generated as in alternative 2 by meansof digital spectrum inversion. Instead of using a low pass filter afterthe DAC in step 3, a band pass filter would be utilized in order toselect the frequencies in the second Nyquist zone in the frequency range[fs/2, fs]. The non-linearity distortions caused by the mixer could beavoided. Furthermore, LO phase noise is circumvented. A possibledisadvantage of this variant might be a higher loss in amplitude atfrequencies close to fs due to the sinc roll-off. This could be avoidedby using return-to-zero (RZ) instead of non-return-to-zero (NRZ)operation for the DAC.

Moreover, in alternative 1 the LO is located in-band and thus maydisturb the signal (waveform). In order to reduce this interference, a(e.g. very good) notch filter could be employed to cancel the LO line.However, this might produce degradations of a time domain waveform.Though, a frequency domain waveform, e.g. OFDM, might be affected less.Furthermore, if the phase of the LO was known, a digital LO could begenerated and inserted in the digital signal, which cancels thedisturbing LO line in the analog signal.

The block diagram shown in FIG. 4 relates the spectra shown in FIG. 3 tothe system 1 depicted in FIG. 1. It is noted that the processing unit 21and the digital signal processor 22 (and thus the pre-equalizing) arecombined in a common unit.

FIG. 5 illustrates the spectral representation S of the received signalbeing restricted to the frequency range [−fs, fs], corresponding to asampling rate of 2fs. Although the received signal is a real valuedsignal, a two-sided spectrum is shown here instead of the single-sidedspectrum, because the DFT yields a two-sided spectrum. The spectrum issplit into a lower frequency band S_(I)(k) and an upper frequency bandS_(II)(k), as already set forth above:S=[S ⁺ ,S ⁻]=[S _(I) ⁺ ,S _(II) ⁺ ,S _(II) ⁻ ,S _(I) ⁻],S _(I)=[S _(I) ⁺ ,S _(I) ⁻]S _(II)=[S _(II) ⁺ ,S _(II) ⁻]

Another representation of the spectra S_(I)(k) and S_(II)(k) is shown inFIG. 6. The spectra S_(I)(k) and S_(II)(k) are each separated into thedesired main components and the undesired cross talk components. Asalready discussed above, the problem encompasses cross coupling termswhich are the mirror spectra of the actual spectra. The arrows in FIG.6, which denote the direction of frequency, visualize that the spectraare reversed in frequency and complex conjugated. This operation can bedescribed by a shift of half of the sample points of the discreteFourier transformation (DFT): X(k+N/2)=X(k−N/2), thereby exploiting therepetitive nature of the discrete spectrum.

As already set forth above, the cross coupling problem, which arises dueto the non-ideal filtering as shown in FIG. 5, is denoted asS _(I)(k)=H ₁₁(k)·X ₁(k)+H ₁₂(k)·X ₂(k+N/2)S _(II)(k)=H ₂₁(k)·X ₁(k+N/2)+H ₂₂(k)·X ₂(k),where X₁(k) and X₂(k) are the spectra after the equalizer (afterperforming the pre-equalizing of the first and the second signal). ThisMIMO system is visualized in FIG. 7. Straightaway, the difference to astandard MIMO problem becomes visible with two additional shiftoperations in the frequency domain. The derived model is a special 2×2MIMO model and can also be rewritten as a standard 4×4 MIMO model asdescribed above. Other approaches such as a 1×1, a 2×1 or even a 4×1model are possible as well.

FIG. 8 depicts a signal processing system 1 according to anotherembodiment of the invention. System 1 comprises a first, a second and athird DAC 31, 32, 33 receiving a digital (sampled) input signal from aDSP unit 22. A processing unit 21 for splitting the sampled input signalinto split signals corresponding to different frequency portions of thesampled signal is provided by the DSP unit 22 or might be realized by aseparate unit.

The processing unit 21 splits the input signal into a first and a secondsignal corresponding to a first and a second frequency portion of thesampled signal, wherein the first signal is transmitted to the first DAC31. The second signal is split into a first and a second subsignal,wherein the first subsignal is supplied to the second DAC 32, while thesecond subsignal is supplied to the third DAC 33. The first subsignalcorresponds to the real part of the second signal and the secondsubsignal corresponds to the imaginary part of the second signal (seeFIG. 10).

The processing system 1 further comprises an IQ mixer 600 (comprising alocal oscillator 601) receiving an analog output signal of the secondDAC 32 and an analog output signal of the third DAC 33. The IQ mixer 600mixes the output signals of the DACs 32, 33 and transmits its output toa combiner 4 (via an analog filter 54). The combiner 4 thus combines theoutput signal of the IQ mixer 600 with the output signal of the firstDAC 31. It is noted that the output signals of the DACs 31-33 are fed tothe combiner 4 and the IQ mixer 600, respectively, via analog filters51-53. However, at least some of the filters 51-54 may be omitted asindicated in FIG. 9.

Due to the IQ mixer 600, the spectrum of the second signal does not needto possess conjugate symmetry properties corresponding to a real valuedtime domain signal. Thus, the spectrum can be defined for the positiveas well as for the negative frequencies independently and the resultingtime domain signal is complex valued. FIG. 10 shows the partitioning ofthe input signal (data) spectrum. The spectrum is split into two parts.The first part (comprising the spectrum portion I⁻, I⁺ corresponds to areal valued signal which is fed directly to the first DAC 31. The secondpart of the spectrum (comprising the spectrum portion II⁺, III⁺) doesnot show conjugate symmetry such that the corresponding time domainsignal is complex. After a Fourier transformation (e.g. an IFFT), thetime domain signal is separated into the real and the imaginary part(i.e. the first and the second subsignal), which are fed to the secondand the third DAC, respectively.

As mentioned above, splitting of the non-conjugate-symmetrical spectrumfor the first and the second DAC can be performed in the spectral domaininstead of the time domain as well. Therefore, the individual signalsfor the inphase component (second DAC 32) and the quadrature component(third DAC 33) can be obtained each by a Fourier transformation (e.g. anIFFT) of certain spectral components directly. This step requires theexploitation of the general symmetry properties of the Fouriertransformation for odd and even functions and spectra, respectively.

It is further noted that more than three DACs might be used and that theDACs 31-33 of FIGS. 8 and 9 do not need to be separate standard singleDACs. For example, at least some of them might be realized by aplurality of sub-DACs. For example, the sub-DACs 300 a-300 k can beeither implemented as TIDACs using digital time interleaving incombination with an analog summation point 7 as shown in FIG. 11a ).Further, they might be realized as MUXDACs with an analog multiplexer 70(MUXDAC) as shown in FIG. 11b ). The MUXDAC realization does notnecessarily need to combine all outputs of the DACs 300 a-300 k in asingle multiplexer. Rather, a multiple-stage multiplexer is possible aswell, e.g. in the case of 8 DACs: 3 stages of 2:1 multiplexers (firststage: 4×2:1, second stage: 2×2:1, third stage: 1×2:1. Note, that theindividual sub-DACs 300 a-300 k could be frequency interleaved as well,thereby creating multiple hierarchies of frequency interleaved DACs.

Further, the system 1 may also comprise a pre-equalizing unit 220 asdiscussed above with respect to FIGS. 1 and 2, wherein thepre-equalizing unit is configured for pre-equalizing the signalssupplied to DACs 31-33.

FIG. 12 illustrates another processing system 1 according to theinvention based on the system shown in FIG. 1 and configured forcarrying out a calibration procedure. In order to carry out acalibration, the system 1 comprises a channel identification unit 80used for carrying out a channel estimation regarding the first and thesecond processing channel 101, 102. A splitter 81 is used for branchingoff a portion of the output signal of the combiner 4, the branched offportion being supplied to the channel identification unit 80. Of course,a switch could be used instead of the splitter as well.

FIGS. 13a ) to 13 c) show different realizations of the channelidentification unit 80. According to FIG. 13a ), the full spectrum ofthe signal (produced by splitter 81) is acquired (using an oscilloscope800) and processed by a DSP/CE block 801. According to FIG. 13b ),fractions of the signal are obtained by filtering (using a filter 802)and down conversion (using a mixer 803), thus limiting the requirementsfor the bandwidth and the sample rate of oscilloscope 800. The filterand the LO 8031 of the mixer 803 might be tunable, thereby enabling theiterative identification of the full spectrum. The sequentially obtainedinformation is recombined digitally, thus obtaining information aboutthe full spectral width.

Further, in FIG. 13c ) information about the LO's frequency f_(LO) andphase ϕ_(LO) is acquired, only, since the LO 61, 601 is the onlycritical dynamical element in the system setup. It is either obtained bya notch filter in combination with a scope or by reading the informationabout the current frequency and phase from a PLL (using oscilloscope800) for the stabilization of the LO. An initial channel identificationmight be needed for this variant.

The invention is of course not limited to the realizations shown inFIGS. 13a ) to c). For example, hybrid variants as well as relatedvariants are possible. Further, dedicated sequences for the channelestimation are not absolutely necessary, but might improve the channelestimation quality.

Further, as already set forth above, in order to compensate analogimpairments of the mixer 6, the filters 51-53, the combiner 4 and/or thefrequency response of DACs 31, 32, information about the impulseresponses and/or frequency responses of these systems is needed. Thecalibration routine uses a channel estimation algorithm (see above) toretrieve this information for the whole system. However, it is possibleas well to use S-parameter analyzers or X-parameter analyzers to obtainthis information. The system can be either measured as a whole or thecomponent's parameters are measured individually and are digitallycombined afterwards. During operation system 1 might need to compensatefor changing parameters, e.g. component's temperature variations etc. Inparticular, the calibration procedure is used during operation of system1 in order to constantly adapt the system 1. Possible calibrationprocedure have been already explained above.

For example, the calibration procedure uses channel estimation schemesillustrated in FIGS. 14a ), 14 b) and 16. According to FIG. 14a ), onlythe first DAC 31 is running, wherein the signal s′(t) (from splitter 81)is acquired by the oscilloscope 800 of the channel identification unit80. A FFT is computed (using the DSP unit 801) and the resulting signalis down-sampled in the spectral domain to 2f_(s). After splitting thespectrum into two parts, the spectra are converted into the time domain,where two individual Least-Squares (LS) CEs are performed. One of themwith the sequence representing the image spectrum and the other with thesequence representing the regular spectrum. Lastly, the obtained channelimpulse responses are transformed to the frequency domain yieldingH₁₁(k) and H₂₁(k). The same procedure is used to calculate H₁₂(k) andH₂₂(k), but for the difference that only the second DAC 32 is running(FIG. 14b ). The channel frequency responses are later interpolated tomatch the length of the data pattern in order to perform frequencydomain equalization (FDE).

Besides, the CE can be performed with the ABI sequences (i.e. thepayload sequences) as well, but the quality might be improved by usinge.g. a De Bruijn Binary Sequence (DBBS) pattern of equal length. Note,that the channel estimation can be performed in the frequency domain aswell.

In order to circumvent problems with the aforementioned CE scheme,another scheme is presented in FIG. 15. According this scheme, a reducednumber of FFT/IFFT operations is utilized. The CE sequences of thescheme presented in the previous section could determine spectralcomponents in the range [0, f_(s)/2] and [f_(s)/2, f_(s)]. With thesenew sequences spectral components in the range [0, f_(s)] can beestimated in a single step and thus a more efficient and reliableestimation may be obtained.

Another solution (FIG. 16) uses a MIMO least squares channel estimationscheme, whereby the MIMO 2×1 channel is estimated jointly. The resultingfrequency responses are split in the frequency domain in order toretrieve four channel frequency responses.

The DSP steps for the ABI scheme using a repetitive data sequence, e.g.for an arbitrary waveform generator (AWG), are shown in FIG. 17. Using afrequency domain modulation format, e.g. OFDM or DMT, obviates the needfor the first FFT Operation. The input signal (sequence) d(n) istransformed to the frequency domain using a fast Fourier transform(FFT). Then, it is split into two parts D₁(k) and D₂(k). Further, ashifted and a non-shifted version of both D₁(k) and D₂(k) are generated.The equalizers are applied (i.e. the pre-equalizing is performed) to thespectra D_(j)(k) and D_(j)(k+N/2) with i, j∈{1,2}. The resulting spectraare transformed to the time domain via an inverse fast Fourier transform(IFFT) operation and the resulting sequences can be fed to the DACs 31,32. The equalizer coefficients W_(ij)(k) with i, j∈{2, 2} are given bythe solution for the 2×2-MIMO problem as discussed above.

FIG. 18 generally illustrates an example of carrying out the splittingof the input signal and the pre-equalization (which e.g. be used withthe system of FIGS. 1 and 2, but also with the IQ mixer systemillustrated in FIG. 8 or 9). The input sequence d(n) is transformed intoits spectral representation using an FFT. In the spectral domain thesequence is split into two parts (using the processing unit 21), i.e.the first and the second signal are generated. The first partcorresponds to low frequency (LF) components and the second part to highfrequency (HF) components of the input signal. The splitting can be doneby partitioning the spectrum, whereby low frequency samples and highfrequency samples are picked from the frequency domain representation inorder to generate spectral representations of the two sequences (thefirst and the second signal). This operation might comprise a ratechange. In FIG. 18, the rate change is denoted by the downward arrows.

Now, a MIMO equalizer 221 of the pre-equalizing unit 220 follows, whichcompensates (as already mentioned above) e.g.

a) magnitude and/or phase in each frequency band

b) magnitude and/or phase mismatches between the frequency bands and

c) cross talk between the frequency bands.

d) and might also account for non-linear distortions

There are multiple ways of achieving the spectrum split (i.e. forconfiguring the processing unit 21). In the following two possibilitiesare explained. The main condition for the splitting functions is toequal 1 over all discrete frequencies and/or to ensure that all discretefrequency components are present in one or the other frequency portion.

For example (see steps 1 and 2 of above FIG. 3), brickwall filteringmight be used for the lower frequency portion of the input signalcorresponding to ideal low pass filtering combined with down sampling ofe.g. a factor 2 (if the input signal (spectrum) is split equally). Forthe higher frequency portion this operation corresponds to bandpass(highpass) filtering followed by downmixing and an additional ideallowpass filtering action. For example, factor 2 downsampling follows (ifthe input signal is split equally). In the frequency domain this can beachieved by selecting the appropriate samples.

Another possibility is raised cosine filtering (see FIG. 19). Thefrequency samples which are multiplied by zeros in the raised cosinefunction may be removed as illustrated in the figure in order to performa rate conversion. The frequency response of raised cosine filtersH_(rc, low) and H_(rc, high) are illustrated in FIG. 20a ). FIG. 20b )shows the frequency response of brick wall filters H_(block, low) andH_(block, high).

FIG. 21 illustrates a method for data processing according to anothervariant of the invention. Again, at least a first and a second DAC isprovided and a sampled signal is split into a first and a second signalcorresponding to different frequency portions of the sampled signalusing a processing unit (steps I, II). Further, a first and a secondanalog signal is created using the first and the second DAC (step III).However, different from the method shown in FIG. 3, oversampled firstand second signals are generated and converted by the first and thesecond DAC in order to obtain the first and second analog signal. Theoversampling can of course be combined with the pre-equalizing of thesignals described above.

The oversampling (e.g. by inserting zeros in the digital spectra) isused in order to move the image bands away from the desired bands. Thus,appropriate analog filters (step IV and step VI) are able to eliminatethe images almost completely such that cross talk between the processingchannels may be avoided. The input spectrum may be divided unequallyamong the first and the second DAC since the signal in the firstprocessing path undergoes filtering only once and thus does not requireoversampling both at the high frequencies and the low frequencies. Theoversampled second signal is not generated at base band, but at anintermediate frequency (digital upmixing or digital upconversion). Thus,spectral zeros are achieved both at high frequencies and at lowfrequencies. Then, the second signal is upconverted to the desiredfrequency using an LO (step V) and a sideband rejection filter (e.g.band pass, high pass or low pass filter) removes the undesired side band(step VI).

The principle design of a system 1 (being e.g. identical to the systemillustrated in FIG. 1) for carrying out the oversampling methoddescribed above is shown in FIG. 22, wherein the signal spectra atspecific location of the system 1 are also shown.

The invention claimed is:
 1. A signal processing system, comprising atleast a first, a second and a third digital-to-analog converter (DAC); aprocessing unit configured for splitting a sampled signal into at leasta first and a second signal corresponding to different frequencyportions of the sampled signal, transmitting the first signal to thefirst DAC, splitting the second signal into a first and a second subsignal and transmitting the first sub signal to the second DAC and thesecond subsignal to the third DAC, the first subsignal corresponding tothe real part of the second signal and the second sub signalcorresponding to the imaginary part of the second signal; an IQ mixerconfigured for mixing an analog output signal of the second DAC and ananalog output signal of the third DAC; a combiner for combining ananalog output signal of the first DAC and an output signal of the IQmixer.
 2. The system as claimed in claim 1, wherein the frequencyportion that corresponds to the first signal comprises lower frequenciesthan the frequency portion that corresponds to the second signal.
 3. Thesystem as claimed in claim 1, wherein the processing unit is configuredfor carrying out the splitting of the sampled signal into the first andthe second signal in the frequency domain.
 4. The system as claimed inclaim 1, wherein the processing unit is configured for carrying out thesplitting of the sampled signal into the first and the second signal inthe time domain.
 5. The system as claimed in claim 1, wherein theprocessing unit is configured for carrying out a Fourier transform ofthe second signal for generating the first and the second subsignal. 6.The system as claimed in claim 1, further comprising at least one lowpass filter for filtering the outputs of the DACs and/or a band pass ora low pass or a high pass filter for filtering the output of the IQmixer.
 7. The system as claimed in claim 1, wherein the processing unitis realized by a digital signal processor.
 8. The system as claimed inclaim 1, wherein the IQ mixer is configured for single sidebandmodulation.
 9. The system as claimed in claim 1, wherein the IQ mixer isrealized by an opto-electronic modulator.
 10. A signal processingmethod, in particular using the system according to claim 1, the methodcomprising the steps of: providing at least a first and a seconddigital-to-analog converter (DAC); splitting a sampled signal into atleast a first and a second signal corresponding to different frequencyportions of the sampled signal by means of a processing unit;pre-equalizing the first and the second signal; converting thepre-equalized first signal into a first analog signal using the firstDAC; converting the pre-equalized second signal into a second analogsignal using the second DAC; combining the first and the second analogsignal using a combiner, wherein the processing unit, the first DAC andthe combiner define a first processing channel, wherein the processingunit, the second DAC and the combiner define a second processingchannel, wherein the pre-equalized first signal is generated byprocessing the first signal in such a way that the pre-equalized firstsignal compensates cross talk between the first and the secondprocessing channel, and/or the pre-equalized second signal is generatedby processing the second signal in such a way that the pre-equalizedsecond signal compensates cross talk between the first and the secondprocessing channel.
 11. The method as claimed in claim 10, whereingenerating the pre-equalized first and second signal is carried outusing the results of a calibration measurement with respect to at leasta spatial, frequency and/or time portion of the first and/or the secondprocessing channel.
 12. The method as claimed in claim 11, wherein thecalibration measurement is carried out using a channel estimation schemewith respect to the first and/or the second processing channel.
 13. Themethod as claimed in claim 12, wherein the channel estimation schemecomprises treating the combination of the first and the secondprocessing channel as a MIMO system.
 14. The method as claimed in claim13, wherein the calibration measurement comprises determiningcoefficients of a frequency response matrix related to the MIMO system.15. The method as claimed in claim 12, wherein the channel estimationscheme comprises transmitting a channel estimation sequence to the firstand/or the second DAC.
 16. The method as claimed in claim 15, wherein afirst channel estimation sequence is transmitted to the first DAC and asecond channel estimation sequence is transmitted to the second DAC,wherein the first channel estimation sequence is distinguishable fromthe second channel estimation sequence.
 17. The method as claimed inclaim 11, wherein the calibration measurement comprises an S- and/orX-parameter measurement of at least a part of an analog section of thefirst and/or the second processing channel.
 18. The method as claimed inclaim 10, wherein the pre-equalized first and second signal aregenerated adaptively by means of the results of re-calibrationmeasurements carried out using a portion of an analog signal produced bythe combiner.
 19. The method as claimed in claim 10, further comprisingcreating an oversampled first signal and converting the oversampledfirst signal by the first DAC in order to obtain the first analog signaland/or creating an oversampled second signal and converting theoversampled second signal by the second DAC in order to obtain thesecond analog signal.
 20. A signal processing system, in particular forcarrying out the method according to claim 10, the system comprising: aprocessing unit configured for splitting a sampled signal into at leasta first and a second signal corresponding to different frequencyportions of the sampled signal; a pre-equalizing unit for pre-equalizingthe first and the second signal; at least a first digital-to-analogconverter (DAC) for converting the pre-equalized first signal into afirst analog signal and a second DAC for converting the pre-equalizedsecond signal into a second analog signal; a combiner for combining thefirst and the second analog signal, wherein the processing unit, thefirst DAC and the combiner define a first processing channel, whereinthe processing unit, the second DAC and the combiner define a secondprocessing channel, wherein pre-equalizing unit is configured forgenerating the pre-equalized first signal by processing the first signalin such a way that the pre-equalized first signal compensates cross talkbetween the first and the second processing channel, and/or forgenerating the pre-equalized second signal by processing the secondsignal in such a way that the pre-equalized second signal compensatescross talk between the first and the second processing channel.